N-path mixer-based receiver apparatus and method with transmitter self-interference rejection

ABSTRACT

An apparatus comprising: a receiver; and one or more N-path filters coupled to an input of the receiver, wherein the one or more N-path filters apply a combination of non-overlapping pulses and a pseudo noise (PN) code.

CLAIM FOR PRIORITY

This application claims priority to U.S. Provisional Patent ApplicationNo. 62/505,772, filed on 12 May 2017, titled “N-PATH MIXER-BASEDRECEIVER WITH TRANSMITTER SELF-INTERFERENCE REJECTION,” and which isincorporated by reference in its entirety.

GOVERNMENT SUPPORT STATEMENT

The embodiments of the invention were made with the support of theUnited States Government under Award No. N66001-14-1-4013 awarded by theDefense Advanced Research Projects Agency (DARPA). The Government hascertain rights in the invention.

BACKGROUND

Emerging 5G applications rely on increased spectrum reuse to increasenetwork capacity, leading to increase in in-band or co-channelinterference (CCI) in receivers. While phased-arrays (e.g.,Multiple-input-Multiple-output (MIMO) arrays) enable spatial filteringof CCI, digital beamforming (DBF), in such arrays it is desirable forreconfigurable, concurrent multiple beams. However, the absence ofanalog spatial filtering results in high Analog-to-Digital (ADC) dynamicrange requirements to tolerate CCI/jammers. The absence of analogspatial filtering has led to interest in development of notched-arrayswith spatio-spectral notching of jammers/interferers in RF/analog priorto the ADC and DBF. Prior spatio-spectral notching approaches includespatio-spectral filtering of one blocker at an RF (radio frequency)input in an X-band array and spatio-spectral filtering of one blockerusing feed-forward cancellation (FFC) at the baseband.

BRIEF DESCRIPTION OF THE DRAWINGS

The embodiments of the disclosure will be understood more fully from thedetailed description given below and from the accompanying drawings ofvarious embodiments of the disclosure, which, however, should not betaken to limit the disclosure to the specific embodiments, but are forexplanation and understanding only.

FIG. 1 illustrates an apparatus with spatio-spectral filtering at an RFinput to attenuate multiple blockers (both out-of-band and in-band atspecific angles of incidence) which reduces block levels at an input ofan analog-to-digital converter (ADC) enabling subsequent digitalbeamforming, according to some embodiments of the disclosure.

FIGS. 2A-B illustrate a code-domain receiver implementation using N-pathfilters, and associated control signals and apparatus, in accordancewith some embodiments.

FIG. 3 illustrates a frequency-domain/code-domain notch filterimplemented using an N-path filter, in accordance with some embodiments.

FIG. 4 illustrates a frequency-domain/code-domain parallel notch filterimplemented using an N-path filter, in accordance with some embodiments.

FIG. 5 illustrates a reconfigurable RX architecture for parallelspatio-spectral notch filtering (PSNF), in accordance with someembodiments.

FIG. 6A illustrates 3^(rd) order and two 2^(nd) order Walsh functionsequences (WF-seq), according to some embodiments.

FIGS. 6B-C illustrate schematics of a four-element PSNF with fourcorrelators in each element operating from a low frequency (e.g., 0.3GHz) to a high frequency (e.g., 1.4 GHz), in accordance with someembodiments.

FIG. 6D illustrates waveforms of 3^(rd) order and two 2^(nd) order Walshfunction sequences, according to some embodiments.

FIGS. 7A-B illustrate plots showing measured S11 parameters for one PSNFinput/output pair with 3^(rd) order WF-seq for single frequency and two2^(nd)-order WF-seq for concurrent dual-frequency tunable notch,respectively, in accordance with some embodiments.

FIGS. 8A-B illustrate plots showing measured S21 parameters for one PSNFinput/output pair with 3^(rd) order WF-seq for single frequency and two2^(nd)-order WF-seq for concurrent dual-frequency tunable notch,respectively, in accordance with some embodiments.

FIGS. 8C-D illustrate measured array gain for a 4-element array for twosettings demonstrating concurrent dual frequency/AoI(angle-of-incidence) notch filtering, in accordance with someembodiments.

FIG. 8E illustrates measured constellation with and without notchfiltering for wireless test setup with in-band −10 dBm AWGN (Additivewhite Gaussian noise) blocker at the RF (Radio Frequency) input, inaccordance with some embodiments.

FIG. 9 illustrates a die photo of an integrated circuit (IC) having a4-element PSNF and performance summary, in accordance with someembodiments.

FIG. 10 illustrates an N-path mixer based receiver (RX) withnon-overlapping pulse local oscillator (LO), in accordance with someembodiments.

FIG. 11 illustrates a 3^(rd) order Walsh-function sequence (WF-seq) fora transmitter (TX) and a receiver (RX), in accordance with someembodiments.

FIG. 12 illustrates a plot showing the SI at N-path RX output followingde-spreading using PN code and WF-seq/PN code, in accordance with someembodiments.

FIGS. 13A-C illustrate schematics of a dual-correlator gain-boostedN-path RX, according to some embodiments of the disclosure.

FIG. 14A illustrates a die photo of an integrated circuit (IC) of 65 nmCMOS N-path RX, in accordance with some embodiments.

FIG. 14B illustrates a plot showing measured S11 parameter when W_(CH1)and W_(CH2) in FIG. 13A are 4-phase NOP pulses (without code modulation)at two frequencies, f1 and f2.

FIG. 14C illustrates a plot showing measured available and reflectedpower for matched and mismatched de-spreading codes, in accordance withsome embodiments.

FIG. 15A illustrates a plot showing measured RX gain and isolationbetween a first channel and a second channel, in accordance with someembodiments.

FIG. 15B illustrates a plot showing measured two-tone SI at RX outputfollowing spreading and de-spreading, in accordance with someembodiments.

FIG. 15C illustrates a plot showing measured integrated in-band RXoutput power for modulated SI, in accordance with some embodiments.

FIG. 15D illustrates a plot showing concurrent reception of twocode-modulated RX signals at 750 MHz.

FIG. 15E illustrates a plot showing reception of desired 750 MHz RXsignal in the presence of in-band SI following rejection approach, inaccordance with some embodiments.

FIG. 16 illustrates a flowchart of a method of filtering, in accordancewith some embodiments.

DETAILED DESCRIPTION

Some embodiments provide blocker suppression at a radio frequency (RF)or intermediate frequency (IF) input to address intermodal productsbetween blockers as well as between blockers and a desired signal. Someembodiments describe a scalable, reconfigurable receiver (RX)architecture for parallel spatio-spectral notch filtering (PSNF) thatallows concurrent rejection of two or more blockers at two or moreindependent frequencies/angle-of-incidence (AoI) at each antenna inputin a DBF array. In some embodiments, the PSNF combines orthogonality ofWalsh function sequences (WF-seq) and impedance translation of passivemixers to enable re-configurability, providing high spatio-spectralattenuation (e.g., greater than 20-dB spatio-spectral attenuation) atthe RF input (e.g., 18 dB higher at RF input than state-of-art schemes)for one frequency/AoI notch and higher rejection (e.g., approximately−15 dB rejection) at two independent frequencies/AoI. The approach ofvarious embodiments can be extended to higher frequencies withtechnology scaling. In some embodiments, the approach can be combinedwith feedforward cancellation of jammers/interferers forinterferer-mitigation in Multiple-Input-Multiple-Output (MIMO) arrays.

Some embodiments extend frequency-domain and spatial-domain N-pathfiltering to code-domain filtering by considering sequence mixing usingN-path passive mixers. Some embodiments describe a code-domain N-pathreceiver (RX) based on pseudo noise (PN) code (PN-code) modulated LO(local oscillator) pulses for concurrent reception of two code-modulatedsignals. Some embodiments use a combination of Walsh-Function and PNsequence to translate in-band transmitter (TX) self-interference (SI) toout-of-band at N-path RX output enabling frequency filtering for high SIrejection. In one example, a 0.3 GHz-1.4 GHz 65-nm CMOS (ComplementaryMetal Oxide Semiconductor) implementation is described with 35 dB gainfor desired signals which concurrently receives two RX signals whilerejecting mismatched spreading codes at the RF input. The TX SImitigation approach of some embodiments results in high rejection at theRX input (e.g., 38.5 dB rejection for −11.8 dBm 1.46 Mb/s QPSK(Quadrature Phase Shift Keying) modulated SI at the RX input). In oneexample, the RX achieves 23.7 dBm OP1 dB (gain compression) for in-bandSI, while consuming approximately 35 mW and occupies 0.31 mm². Othertechnical effects will be evident from the various embodiments andfigures.

The embodiments of the disclosure will be understood more fully from thedetailed description given below and from the accompanying drawings ofvarious embodiments of the disclosure, which, however, should not betaken to limit the disclosure to the specific embodiments, but are forexplanation and understanding only.

In the following description, numerous details are discussed to providea more thorough explanation of embodiments of the present disclosure. Itwill be apparent, however, to one skilled in the art, that embodimentsof the present disclosure may be practiced without these specificdetails. In other instances, well-known structures and devices are shownin block diagram form, rather than in detail, in order to avoidobscuring embodiments of the present disclosure.

Note that in the corresponding drawings of the embodiments, signals arerepresented with lines. Some lines may be thicker, to indicate moreconstituent signal paths, and/or have arrows at one or more ends, toindicate primary information flow direction. Such indications are notintended to be limiting. Rather, the lines are used in connection withone or more exemplary embodiments to facilitate easier understanding ofa circuit or a logical unit. Any represented signal, as dictated bydesign needs or preferences, may actually comprise one or more signalsthat may travel in either direction and may be implemented with anysuitable type of signal scheme.

Throughout the specification, and in the claims, the term “connected”means a direct connection, such as electrical, mechanical, or magneticconnection between the things that are connected, without anyintermediary devices.

The term “coupled” means a direct or indirect connection, such as adirect electrical, mechanical, or magnetic connection between the thingsthat are connected or an indirect connection, through one or morepassive or active intermediary devices.

The term “adjacent” here generally refers to a position of a thing beingnext to (e.g., immediately next to or close to with one or more thingsbetween them) or adjoining another thing (e.g., abutting it).

The term “circuit” or “module” may refer to one or more passive and/oractive components that are arranged to cooperate with one another toprovide a desired function.

The term “signal” may refer to at least one current signal, voltagesignal, optical, electromagnetic signal, magnetic signal, or data/clocksignal. The meaning of “a,” “an,” and “the” include plural references.The meaning of “in” includes “in” and on.

The term “blocker” here generally refers to any undesired large signalthat is incident on a receiver with the desired signal. Due to the largesignal strength of the blocker, the RF/analog front-end of the receivergenerally produces an undesired distorted signal and this reduces thegain of the desired signal and causes reception degradation. In somecases, the blocker causes the reception to be impossible by thereceiver.

The term “scaling” generally refers to converting a design (schematicand layout) from one process technology to another process technologyand subsequently being reduced in layout area. The term “scaling”generally also refers to downsizing layout and devices within the sametechnology node. The term “scaling” may also refer to adjusting (e.g.,slowing down or speeding up—i.e. scaling down, or scaling uprespectively) of a signal frequency relative to another parameter, forexample, power supply level.

The terms “substantially,” “close,” “approximately,” “near,” and“about,” generally refer to being within +/−10% of a target value. Forexample, unless otherwise specified in the explicit context of theiruse, the terms “substantially equal,” “about equal” and “approximatelyequal” mean that there is no more than incidental variation betweenamong things so described. In the art, such variation is typically nomore than +/−10% of a predetermined target value.

Unless otherwise specified the use of the ordinal adjectives “first,”“second,” and “third,” etc., to describe a common object, merelyindicate that different instances of like objects are being referred to,and are not intended to imply that the objects so described must be in agiven sequence, either temporally, spatially, in ranking or in any othermanner.

For the purposes of the present disclosure, phrases “A and/or B” and “Aor B” mean (A), (B), or (A and B). For the purposes of the presentdisclosure, the phrase “A, B, and/or C” means (A), (B), (C), (A and B),(A and C), (B and C), or (A, B and C).

The s “left,” “right,” “front,” “back,” “top,” “bottom,” “over,”“under,” and the like in the description and in the claims, if any, areused for descriptive purposes and not necessarily for describingpermanent relative positions.

It is pointed out that those elements of the figures having the samereference numbers (or names) as the elements of any other figure canoperate or function in any manner similar to that described, but are notlimited to such.

FIG. 1 illustrates apparatus 100 for spatio-spectral filtering at an RFinput 101 to attenuate multiple blockers (both out-of-band and in-bandat specific angles of incidence) which reduces block levels at an inputof an analog-to-digital converter (ADC) enabling subsequent digitalbeamforming, according to some embodiments of the disclosure. Here,blockers are illustrated as Blocker1 and Blocker2 with light shaded graytext and arrows.

Apparatus 100 comprises multiple antennas 102 ₁-102 _(N), multiplereceivers 103 ₁-103 _(N), and filter 104. Each receiver may include alow noise amplifier (LNA) 103 a, mixer 103 b, and an ADC 103 c.

In some embodiments, each antenna (e.g., 102 ₁) may comprise one or moredirectional or omnidirectional antennas, including monopole antennas,dipole antennas, loop antennas, patch antennas, microstrip antennas,coplanar wave antennas, or other types of antennas suitable fortransmission of Radio Frequency (RF) signals. In some multiple-inputmultiple-output (MIMO) embodiments, antennas 102 ₁-102 _(N) areseparated to take advantage of spatial diversity.

ADCs, such as ADC 103 c, are apparatuses that convert an analog signalcontinuous physical quantities (e.g., voltages) to a digital signal withdigital numbers that represent the amplitude of the physical quantities.Here, an analog signal is any continuous signal for which the timevarying feature (variable) of the signal is a representation of someother time varying quantity, i.e., analogous to another time varyingsignal. Conversely, a digital signal is a physical signal that is arepresentation of a sequence of discrete values (a quantifieddiscrete-time signal), for example of an arbitrary bit stream, or of adigitized (sampled and analog-to-digital converted) analog signal.

Any suitable ADC may be used to implement ADC 103 c. For example, ADC103 c is one of: direct-conversion ADC (for flash ADC), two-step flashADC, successive-approximation ADC (SAR ADC), ramp-compare ADC, WilkinsonADC, integrating ADC, delta-encoded ADC or counter-ramp, pipeline ADC(also called subranging quantizer), sigma-delta ADC (also known as adelta-sigma ADC), time-interleaved ADC, ADC with intermediate FM stage,or time-stretch ADC. For purposes of explaining the various embodiments,ADC 103 c is considered to be flash ADCs.

The PSNF approach of various embodiments is insensitive to mixersequence overlap and hence, it can be scaled to higher number ofblockers with technology scaling. Prior N-path tunable notch filtersutilize N-phase non-overlapping local oscillator (LO) pulses (NOP) todrive mixer switches. Additionally, prior N-path tunable notch filterscreate a high input impedance at notch frequency leading to high blockervoltage swing at the input.

The filter of some embodiments can be used to attenuate two blockers,one at frequency f₁ and another at frequency f₂ with some angle ofincidence. In the presence of any spatial/spectral blocker, the dynamicrange required from ADC 103 c to establish robust communication withdesired signal is high and causes increase in power consumption. Thenotch filter of some embodiments can be used to implement eitherspectral notch filter or spatial notch filter at block frequency/angleof incidence. Due to the action of the notch filter, the blockers seelow impedance at each antenna interfaces in the MIMO (multiple inputmultiple output) receiver. The notch filter of some embodiments alsoattenuates the blockers even before the receiver input IN_(k).

The outputs of all receivers are used to form digital beam to receiversignals from all direction of arrival. In the presence of proposedfilter, the signal arriving from blocker direction is attenuated.Therefore, at the output of ADC, signals are received from all thedirections with equal gain except in the blocker's angle of arrivaldirection and shown as 105 _(1-N) for each ADC output. These outputs canbe used to extract signals from all direction using digital beam formingtechnique in a MIMO receiver.

FIGS. 2A-B illustrate a code-domain receiver implementation using N-pathfilters, and associated control signals and apparatus, in accordancewith some embodiments. The switches (S₀, S₁, . . . S_(N)) are coupled toan Antenna (ANT) and are driven by sequences (W₀, W₁, . . . , W_(N))which are generated by a combination of non-overlapping pulses (P₀, P₁,. . . P_(N)) and pseudo-random (PRN) or pseudo noise (PN) codes,according to some embodiments of the disclosure. Here, T_(s) is theperiod of the non-overlapping pulses. T_(s) defines the period of thecarrier frequency and is set by the frequency of the blocker.

In some embodiments, each non-overlapping pulse (e.g., one of P₀, P₁, .. . P_(N)) is multiplied using a multiplier 223 with PN code (PN_(c)) togenerate switch control output W. For example, non-overlapping pulse P₀is multiplied with PN_(c) to generate W₀ which controls switch S₀. OtherW_(i) can be generated using similar techniques. Here, Block 1 (BLK₁)221 which converts an LO input to non-overlapping pulses (P₀, P₁, . . .P_(N)), in accordance with some embodiments. FIG. 2B also shows Block 2(BLK₂) which receives a Code Clock and generates it using PRN or PNcodes (PN_(c)). The PRN (same as PN) codes themselves can be a part ofwell-known codes such as M-sequences (maximal length sequences), GoldCodes, Kasami codes, Barker codes, etc., in accordance with someembodiments. The switches of the filter drive a combination ofinductors, capacitors (C₀, . . . C_(N)) and resistors (R₀ . . . R_(N)),in accordance with some embodiments.

FIG. 3 illustrates a frequency-domain/code-domain notch filter 300implemented using an N-path filter, in accordance with some embodiments.In some embodiments, filter 300 is connected in series with the receiver(RX) 301. In some embodiments, RX 301 can reject certain non-overlappingpulses or code modulated signals at the RX input. In some embodiments,the switches are driven by sequences (W₀, W₁, . . . , W_(N)) generatedby a combination of non-overlapping pulses and pseudo-random codes asdiscussed with reference to FIGS. 2A-B. In some embodiments, thepseudo-random codes themselves can be a part of well-known codes such asM-sequences (maximal length sequences), Gold Codes, Kasami codes, Barkercodes, etc., in accordance with some embodiments.

FIG. 4 illustrates an apparatus 400 with frequency-domain/code-domainparallel notch filter implemented using an N-path filter, in accordancewith some embodiments. In some embodiments, the notch filter isconfigured as an array. In some embodiments, the array uses Walshfunction sequence mixing and impedance translation of passive mixers. Insome embodiments, the filter of FIG. 4 is connected in parallel with RX301. In some embodiments, the switches (e.g., S₀, S₁, S₂, S₃ throughS_(N), S_(1N), S_(2N), S_(3N)) of the filter drive a combination ofinductors (L₀, . . . L_(N)) and resistors (R₀ . . . R_(N)). RX 301 canreject certain frequency signals or certain code-modulated signals atthe receiver input. In some embodiments, the switches are driven bysequences (W₀, W₁, . . . , W_(N)) generated by Walsh-Functions or acombination of non-overlapping pulses and pseudo-random codes.

Here, Block 1 (BLK₁) 401 converts an LO input (LO_(IN)) toWalsh-Functions (WF₀, WF₁, . . . WF_(N)), while Block 2 (BLK₂) 402receives a Code Clock and generates it using PRN or PN codes (PN_(c)).In some embodiments, each Walsh-Function (e.g., one of WF₀, WF₁, . . .WF_(N)) is multiplied using a multiplier 403 with PN code (PN_(c)) togenerate switch control output W. For example, Walsh-Function WF₀ ismultiplied with PN_(c) to generate W₀ which controls switch S₀. In someembodiments, the pseudo-random codes (e.g., PRN or PN) themselves can bea part of well-known codes such as M-sequences (maximal lengthsequences), Gold Codes, Kasami codes, Barker codes, etc., in accordancewith some embodiments.

FIG. 5 illustrates a reconfigurable RX architecture 500 for parallelspatio-spectral notch filtering (PSNF), in accordance with someembodiments. Architecture 500 comprises antenna models 502 _(1-N)(e.g.,to model antennas 102 _(1-N)), receivers RX_(1-N) 103 a _(1-N), andparallel spatio-spectral notch filter array 104/504. In someembodiments, the antenna models 502 _(1-N) comprise a current sourceproviding I_(ANT) and antenna resistance RANT. In some embodiments,parallel spatio-spectral notch filter array 104/504 comprises sets ofWalsh-Function Correlator Elements 1 (e.g., 504 a ₁-504 a _(N)) throughsets of Walsh-Function Correlator Elements N (504 b _(1-N)).

In some embodiments, in the PSNF approach each set of element comprisesswitches (e.g., S₀₋₃ in element 1 504 a ₁, and S_(N-3N) in element N 504b ₁) which are driven by WF-sequence (e.g., WF signals including WF_(k)and WFb_(k), where WFb_(k) is an inverse of WF_(k)) instead of NOP. Insome embodiments, each set of switches (or at least one set of switches)and baseband gyrator 505 and capacitor C_(BB) can be considered to be acorrelator that senses RF voltage and return current based on theprojection of the RF input voltage on a basis function determined by themixer switches.

In some embodiments, an input signal that leads to low baseband voltageon C_(BB,K) (black as opposed to light gray) leads to small basebandcurrent, I_(BB,K) and small RF current, I_(FILT). This translates to ahigh PSNF input impedance and hence no filtering. However, a blockersignal that is correlated with the mixer switching signals leads to anon-zero voltage on C_(BB,K) (light gray) and hence to gyrator outputcurrents that are up-converted with I_(FILT) following antenna currentI_(ANT). This creates a low RF impedance, attenuating RF input voltage.

Each PSNF element (e.g., 504 a _(1-N)) comprises four correlators inthis implementation, in accordance with some embodiments. Here, Walshfunction (WF) sequences like the Fourier series, are well known as acomplete orthogonal basis system to represent a signal.

In the PSNF approach of FIG. 5, the switches (e.g., S₀₋₃ with referenceto 504 a ₁) are driven by WF sequence instead of NOP. Each set ofswitches and baseband gyrator/capacitor (e.g., 504 a and C_(BB,K)) canbe considered to be a correlator that senses RF voltage and returnscurrent based on the projection of the RF input voltage on a basisfunction determined by the mixer switches. An input signal that leads tolow baseband voltage on capacitor C_(BB;K) leads to small basebandcurrent, I_(BB;K) and small RF current, I_(FILT). This translates to ahigh PSNF input impedance and hence no filtering. However, a blockersignal that is correlated with the mixer switching signals leads to anon-zero voltage on capacitor C_(BB;K) and hence to gyrator outputcurrents that are up-converted with current I_(FILT) following antennacurrent I_(ANT). This creates a low RF impedance, attenuating RF inputvoltage.

Each PSNF element (e.g., Element 1 through Element N) comprises fourcorrelators in this implementation. Notably, Walsh function sequences(WF-seq) like the Fourier series, are well known as a completeorthogonal basis system to represent a signal (sal(i) and cal(i) in FIG.6D). In some embodiments, WF-seq are applied to the switching mixers (asopposed to NOP) for the following reasons: i) similar to N-phase NOP LO,WF-seq orthogonality implies that passive mixers driven by WF-seq can beconnected together at RF without scaling and notch depth is increasedwith higher order WF-seq; ii) unlike NOP, each correlator in FIG. 5 isalways connected to the RF port with the WF-seq approach ensuring acurrent path for the baseband current; iii) since the correlation for asinusoid input with some WF-seq (e.g., wal(0), cal(2), sal(2), sal(4) inFIG. 6D) results in zero output if both have the same period TO,correlation with those sequences is not required; and iv) harmonicproperties of the WF-seq filter are also equivalent to NOP N-path filter

In some embodiments, spatio-spectral filtering can be achieved byconnecting capacitors corresponding to one set of correlators acrosselements. For example, assume that the WFseq are in-phase in allelements, then a blocker with broadside AoI results in constructiveaddition on capacitor C_(BB;K). This leads to currents, I_(BB;K) at allgyrator outputs, causing low RF impedance and blocker notch filtering atall element inputs, in accordance with some embodiments. In someembodiments, if the desired signal AoI results in null voltage oncapacitor C_(BB;K), and hence null gyrator output, all RF inputs see ahigh PSNF impedance, and the desired signal is unaffected at allelements. In some embodiments, the AoI corresponding to notch filteringcan be steered by changing the relative phase of the WF-seq applied tocorrelators in each element.

FIG. 6A illustrates apparatus 600 with 3^(rd) order and two 2^(nd) orderWalsh function sequences, according to some embodiments.

Apparatus 600 comprises switchable gain-boosted N-path RX 601. Forexample, switchable capacitors 601 a ₁ through 601 a _(N) are coupled toamplifier 601 a. In some embodiments, apparatus 600 further comprisessequence generators 602, 603, 606, and 607; four notch filter elements604, 605, 606, and 607, switches S₁₋₈, and Serial peripheral interface(SPI) 610 to provide access to various internal nodes. Here, the RFinputs are Input1, Input2, Input3, and Input4; and RF outputs areOutput1, Output2, Output3, and Output4.

In some embodiments, each RF input is connected to an output of a notchfilter and then it is available as RF output. In some embodiments, thebaseband output of the filter can be shorted to create a spatial filterand can be disconnected to form a spectral notch filter. In someembodiments, the baseband output can be routed to off-chip to a printedcircuit board (PCB), for example, and can be connected with notchfilters on another die to realize a larger array to form a spatial notchwith narrower beam width.

In some embodiments, the filter is designed to be highly configurablewith its ability to program notch filter frequency, notch direction,filter order, bandwidth of the notch, etc. to optimize systemperformance. In some embodiments, SPI interface 610 is used to controlall digital control signal including loading WF-seq to sequencygenerators (e.g., 602, 603, 608, and/or 609).

In some embodiments, two clocks LO1 and LO2 are used for two notches attwo different frequencies. In some embodiments, Sequency gen block(e.g., 602, 603, 608, and/or 609) uses these clocks to generate therequired WF-seq with appropriate frequency and phase shift to form aspatial notch. In some embodiments, the Sequency gen block (e.g., 602,603, 608, and/or 609) can be programmed to use either clocks to formsimultaneous two frequency notch filter or one frequency notch filter.

FIGS. 6B-C illustrate schematics 620 and 630, respectively, of afour-element PSNF with four correlators 621 ₁-621 ₄ (but can be ‘N’numbers of correlators) in each element operating from a low frequency(e.g., 0.3 GHz) to a high frequency (e.g., 1.4 GHz), in accordance withsome embodiments. In some embodiments, schematic 620 comprisescapacitors C1 and C2, circuits 621 a/b and switches S₀-S₇ controlled byWalsh Function sequences WF_(K) and WFb_(k).

In some embodiments, capacitor C1 is used as a DC (direct current) blockto isolate the DC bias of notch filter from the RF node. In someembodiments, capacitor C2 and two trans-conductors are used to implementeffective inductance of the notch filter thus can be used to program thebandwidth. In some embodiments, the input to capacitor C1 is an RF inputsignal which is also routed as RF output in FIG. 6A. In someembodiments, the value of capacitor C1 is large as it is used as DCblock.

In some embodiments, schematic 630 comprises Walsh Function sequencegenerators 631 ₁-631 ₄ (but can be ‘N’ numbers of correlators), whereeach generator receives LO1 and LO2 signals and generates a WalshFunction sequence WF_(k) using a chain of sequential units. In someembodiments, the sequential units are flip-flops 633 ₁ through 633 _(N)that receive clock input from one of LO1 or LO2 provided by multiplexer632.

In some embodiments, the SEL signal is controlled through the SPIinterface 610 and is used to select a desired LO signal to be used tocreate a notch filter. In some embodiments, Walsh Seq. is loadedparallel to the sequency generator using the SPI interface 210 through acomputer interface. In some embodiments, the Walsh Seq. is loaded to aflip-flop using set and reset signals to the flip-flop. In someembodiments, the flip-flops are coupled together in a ring, such thatthe output Q of flip-flop 633 _(N) to in the data input D of the firstflip-flop 633 ₁. The outputs WF_(k) from each generator (e.g., fromgenerator 631 a) is provided to schematic 620 to control the switches.

FIG. 6D illustrates waveforms 640 of 3^(rd) order and two 2^(nd) orderWalsh function sequences, according to some embodiments. Similar toFourier signal that consists of sinusoid and co-sinusoid signals, theWalsh function family consists of infinite series of binary valuewaveform. The first few signals of this family consists of infinitenumber of functions and can be denoted as wal, sal, and cal. Thesesignals are produced at sequency generator output and used as clockdriving switches in the notch filter, in accordance with someembodiments.

In some embodiments, WF-seq are applied to the switching mixers (asopposed to NOP) for the following reasons: i) similar to N-phase NOP LO,WF-seq orthogonality implies that passive mixers driven by WF-seq can beconnected together at the RF without scaling and notch depth isincreased with higher order WF-seq; ii) unlike NOP, each correlator inFIG. 5 is connected to the RF port with the WF-seq approach ensuring acurrent path for the baseband current; iii) since the correlation for asinusoid input with some WF-seq (wal(0), cal(2), sal(2), sal(4)—gray inFIG. 6D) results in zero output if both have the same period To,correlation with those sequences is not required; and iv) harmonicproperties of the WF-seq filter are also equivalent to NOP N-pathfilters.

In some embodiments, spatio-spectral filtering can be achieved byconnecting capacitors corresponding to one set of correlators acrosselements (FIG. 5). For example, when the WF-seq are in-phase in allelements, a blocker with broadside AoI results in constructive additionon C_(BB,K) (FIG. 5). This leads to currents, I_(BB,K) at all gyratoroutputs (e.g., outputs of 505), causing low RF impedance and blockernotch filtering at all element inputs. If the desired signal AoI resultsin null voltage on C_(BB,K), and hence null gyrator output, all RFinputs see a high PSNF impedance, and the desired signal is unaffectedat all elements. The AoI corresponding to notch filtering can be steeredby changing the relative phase of the WF-seq applied to correlators ineach element. In some embodiments, since the correlators in the PSNFcorrelate input voltage and return current, the approach is not affectedby overlap between WF-seq across correlators. This is unlike prior artN-path filters with NOP that correlate current and return voltage,making them sensitive to overlap.

In some embodiments, the PSNFs insensitivity to overlap between basisfunctions allows arbitrary WF-seq to be applied at each correlator. Forinstance, if single-frequency/AoI 3^(rd)-order WF-seq (FIG. 6D) areapplied to the four correlators in each array element, a singlefrequency/AoI notch filter is created which is equivalent to 8-phase NOPfilter. In some embodiments, a concurrent dual-frequency/AoI notch canbe achieved by applying two 2^(nd)-order WF-seq at two independentfrequencies (FIG. 6D) to the four correlators in each element. In someembodiments, the gyrator capacitors C_(BB,K) in the PSNF approach alsocapture the blocker signal for subsequent FFC.

In some embodiments, a gain-boosted N-path RX is included at one elementoutput to demonstrate an RX following the PSNF. Since a G_(M1) followingthe mixer, as in FIG. 5, leads to high flicker noise or capacitiveswitching losses, a translational approach is adopted with G_(M1)preceding the mixer, in accordance with some embodiments. In someembodiments, the WF-seq applied to each LO are generated using aprogrammable shift register (e.g., flip-flops 633 _(1-N)) where thedesired WF-seq is moved along the dual-edge shift register using a 4fLOclock. In some embodiments, a multiplexer allows two of the correlatorsto be shifted using a second LO (LO2) that is used to define the 2^(nd)notch frequency/AoI. 3-bit phase shift can be achieved by changing therelative phase of the WF-seq in shift registers corresponding todifferent elements. The architecture also supports traditional LO-pathphase shifting for higher phase resolution.

FIGS. 7A-B illustrate plots 700 and 720 showing measured S11 parametersfor one PSNF input/output pair with 3^(rd) order WF-seq for singlefrequency and two 2^(nd)-order WF-seq for concurrent dual-frequencytunable notch, respectively, in accordance with some embodiments. Inthis example, the 4-element PSNF is implemented in a 65-nm CMOS andoperates from 0.3 GHz to 1.4 GHz.

FIGS. 8A-B illustrate plots 800 and 820 showing measured S21 parametersfrom one of the RF inputs to the RF output in FIG. 6A for one PSNFinput/output pair with 3^(rd)-order WF-seq for single frequency and two2^(nd)-order WF-seq for concurrent dual-frequency tunable notch,respectively, in accordance with some embodiments.

FIGS. 8C-D illustrate measured array gains 830 and 840 for a 4-elementarray for two settings demonstrating concurrent dual frequency/AoI notchfiltering, respectively, in accordance with some embodiments. In thisexample, the PSNF achieves 14 dB and 20 dB spectral notch depth for2^(nd) and 3^(rd) order WF-seq, respectively. Measured 4-element PSNFarray factor at each output is shown across AoI/frequency for twofrequency and relative phase shift settings of WF-seq, demonstratingconcurrent dual-frequency/AoI spatio-spectral notch filtering (See, FIG.8C and FIG. 8D). The gain-boosted N-path RX connected to Element 1 inFIG. 6A measured approximately 42-dB gain and NF of approximately −2.5dB to 3.9 dB. Measured NF across notch frequency demonstrates equivalent1-element NF of 5.3 dB to 7.5 dB. The measured RX OIP3 (output thirdorder intercept point) is approximately 28 dBm with the PSNF enabled andconfigured for dual-frequency notch.

FIG. 8E illustrates measured constellation 850 with and without notchfiltering for wireless test setup with in-band −10 dBm AWGN (additivewhite Gaussian noise) blocker and a −35 dBm 16 QAM (quadrature amplitudemodulation) modulated desired signal at the RF input, in accordance withsome embodiments. The constellation before and after the spatial blockerfiltering in FIG. 8E demonstrates the efficacy of the notch filteringapproach of some embodiments.

FIG. 9 illustrates a die photo 900 of an integrated circuit (IC) havinga 4-element PSNF and performance summary, in accordance with someembodiments. Here, the IC has extensive programmability in basebandgyrators/capacitors in FIG. 6A, which increases area. The core elementarea is 0.48 mm². Additional measurements demonstrating tiling of twoICs together to achieve 8-element PSNF array have also been performeddemonstrating scalable spatio-spectral filtering at RF input in DBFarrays.

FIG. 10 illustrates a code-domain N-path mixer 1000 based receiver (RX)with non-overlapping pulse local oscillator (LO), in accordance withsome embodiments.

In the N-path passive mixer driven by the non-overlapping LO pulses, theswitch and baseband filter correlate input RF current with the LOsignal. The baseband voltage, VC,k (t) is given by:

$\begin{matrix}{{V_{C,K}(t)} = {\frac{1}{R_{A}C}{\int{\left( {V_{RF} - {\sum{V_{C,J}W_{J}}}} \right)W_{K}{dt}}}}} & (1)\end{matrix}$where, R_(A)=R_(ANT)+R_(SW), and where R_(ANT) is the resistance of theantenna, and where R_(SW) is the resistance of the switch in the ONstate. Orthogonal basis functions as switch signals (e.g., NOP pulseswhere W_(K)=ϕ_(K)) result in the baseband capacitors developing voltagescorresponding to the RF input projection on the basis functions. Sinceeach path operates on the RF current, it is desirable to have no overlapbetween signals driving the switches to avoid charge sharing. Therefore,concurrent operation of parallel correlators driven by NOP at differentfrequencies is unfeasible.

Here, the following are assumed: a desired RX signal, V_(RF)(t) withsymbol rate, B_(R) that is spread using a pseudo-noise (PN) code,PN_(R1) with chip rate, C_(R)=1/T_(C), (code length, M=C_(R)/B_(R)) andtranslated to center frequency, f₀. In this case, if the LO signalapplied to each switch is generated by multiplying the NOP with period,T₀=1/f₀, with a pseudo-noise sequence, PN_(LO), W_(K) is given by,

$\begin{matrix}{W_{K} = \left\{ \begin{matrix}{{{\varphi_{K}(t)}\mspace{14mu}{for}\mspace{14mu}{{PN}_{LO}(t)}} = 1} \\{{{\varphi_{K}\left( {t + {\pi/\omega_{0}}} \right)}\mspace{14mu}{for}\mspace{14mu}{{PN}_{LO}(t)}} = {- 1}}\end{matrix} \right.} & (2)\end{matrix}$then the baseband output signal is given by,

$\begin{matrix}{{V_{C,K}(t)} = {\frac{1}{R_{A}C}{\int{\left( {{V_{RF}{PN}_{R\; 1}} - {\sum{V_{C,J}W_{J}}}} \right)W_{K}{dt}}}}} & (3)\end{matrix}$

Therefore, V_(C,K)(t) depends upon cross-correlation between PN_(LO) andPN_(R1). For PN_(LO)=PN_(R1) (assuming synchronization), de-spreadingoccurs with V_(C,K) (t) given by equation (1) (ignoring bandwidthconstraints). In some embodiments, low cross-correlation between PN_(LO)and PN_(R1) leads to rejection. Impedance translation also occurs in thespectral/code-domain with input RF matching for signals at frequency f₀that are spread with PN_(R1) and RF mismatch for PN codes orthogonal tothe PN_(R1). In some embodiments, the PN code family can be selectedbased on auto-correlation (for synchronization) and cross-correlation(for interferer rejection) properties. While interferer rejection can beincreased by using specific codes such as Gold codes, the rejection canbe insufficient for TX SIRC (simple infrared transmitter code). In someembodiments, a technique is provided to select pairs of PN sequencesthat can provide high SIRC by leveraging a combination of PN and Walshsequences.

FIG. 11 illustrates a 3^(rd) order Walsh-function sequence (WF-seq) forself-interference cancellation for a transmitter (TX) and a receiver(RX), in accordance with some embodiments.

In some embodiments, TX Correlator (Corr) 1101 is positioned in thetransmitter, and is used to spread the signal intended for transmission.For example, the signal intended for transmission is spread bymultiplying the transmitter input signal TX_(IN) with the pseudo codePN_(WF,1).

In some embodiments, block 1102 is a circulator which comprises of threeterminals connected to the TX, RX and antenna 102. In some embodiment,block 1102 is used to isolate the transmit signal (e.g., output of block1101) from the received signal RX_(IN). In an ideal circulator, thetransmitted signal from transmitter only appears at the antenna andsignal received at the antenna terminal only appears at receiver. Thus,the circulator provides the required isolation between the transmitterand the receiver for them to operate simultaneously. The practicalcirculator, however, may suffer from limited isolation between its threeterminals and some leakage transmitted signal may appear at receiverinput RX_(IN).

In some embodiments, block 1103 is a receiver correlator (RX Corr) whichis used to de-spread the code modulated received signal in the receiverby multiplying the received signal with a pseudo code PN_(WF,2).

Walsh-function sequences (WF-seq) are orthogonal sequences well suitedfor digital synthesis. Notably, a PN sequence multiplied by a WF-seq isalso a PN sequence. Considering two PN sequences, PN_(WF,1) andPN_(WF,2),PN _(WF,1) =PN ₁ ·sal(1);PN _(WF,2) =PN ₁ ·cal(3)  (4)

where sal(1) and cal(3) are shown in FIG. 11 and PN₁ is a pseudo-noisesequence with chip rate, C=1/T_(C).

Assuming PN_(WF,1) and PN_(WF,2) are used as the spreading sequence forthe TX and the desired RX signals TX_(IN) and RX_(IN), respectively.Hence, if W_(J) applied to the mixers is generated using PN_(WF,2), andassuming that the TX is synchronized to the de-spreading sequencePN_(WF,2), the SI and desired RX at the baseband output are given by,V _(BB,SI) =V _(TX) ·PN _(WF,1) PN _(WF,2) =sal(4)·V _(TX)  (5)V _(BB,RX) =V _(RX) ·PN _(WF,2) ·PN _(WF,2) =V _(RX)  (6)

From equation (5), SI interaction with RX de-spreading code translatesSI outside RX bandwidth, enables frequency-domain filtering in theN-path RX baseband. While equation (5) is analyzed for one combinationof WF-seq, other WF-seq combinations (FIG. 11) lead to the same result,with higher-order WF-seq providing higher number of pairs.

FIG. 12 illustrates a plot 1200 showing the SI at N-path RX outputfollowing de-spreading using PN code and WF-seq/PN code, in accordancewith some embodiments. Plot 1200 shows the SI after spreading andde-spreading for 1.46 Mb/s (TX1) and 5.85 Mb/s QPSK (TX2) signals spreadwith a chip rate of 93.75 Mchips/s (Mcs) using PN_(WF,1).

The simulated baseband SI output following de-spreading with PN_(WF,2)1202 and 1203, and with another PN code, PN_(X) 1201 is shown. ComparingPN_(WF,2) and PN_(X) for TX1, frequency filtering the SI output at theRX in the PN_(WF,2) case leads to higher SIRC (for integration BW: 746KHz, SIRC with PN_(X): 17.7 dB and with PN_(WF,2): 41.7 dB). A higherratio between chip rate and symbol rate also leads to higher SIRC with41.7 dBm SIRC for TX1 (BW: 746 KHz) compared to 29.6 dB for TX2 (BW:5.85 MHz). The synchronization used between the TX and RX spreadingcodes implies that the TX code must also be aligned with the desired RXsequence using pilot signals similar to a CDMA (Code Division MultipleAccess) RX.

FIGS. 13A-C illustrate schematics 1300, 1320, and 1330, respectively, ofa dual-correlator gain-boosted N-path RX (e.g., a code-domain RX),according to some embodiments of the disclosure.

The correlator-based approach of some embodiments is implemented using again-boosted N-path RX which comprises Channel 1 and Channel 2 4-pathfilters 1301 and 1302, respectively. In some embodiments, the 4-pathfilter 1301 comprises a parallel combination of resistor R1 andcapacitor C1, trans-conductor 1301 a, switch S1 controllable byW_(CH1,0-3), and capacitor C2 coupled together as shown. In someembodiments, the 4-path filter 1302 comprises a parallel combination ofresistor R1 and capacitor C1, trans-conductor 1301 b, switch S1controllable by W_(CH2,0-3), and capacitor C2 coupled together as shown.In some embodiments, W_(CH1,0-3) is generated by logic comprisingcircuit 1303 for non-overlap phase generation and divide-by-twocircuitry which receives LO₁. Schematic 1320 illustrates the generationLO₁. In one example, LO₁ is generated by mixing of clock 1 (CLK₁) withPN_(CH,1) by mixer or multiplier 1321. The output of multiplier 1321 isthen compared by XOR 1323 with its delayed version delayed by delaycircuitry 1322.

In some embodiments, W_(CH2,0-3) is generated by logic comprisingcircuit 1304 for non-overlap phase generation and divide-by-twocircuitry which receives LO₂. In some embodiments, LO₂ is generated bycircuitry 1330 similar to schematic 1320 but using CLK₂ instead of CLK₁,and using PN_(CH,2) instead of PN_(CH,1). An SPI interface 1305 isprovided to get access to internal nodes of the gain boosted N-path RX.In some embodiments, the RF input (V_(RF)) is received by an LNA whichis coupled to a feedback resistor R2 such that the input of the LNA isalso coupled to capacitor C2 of 1301 while the output of LNA is coupledto capacitor C2 of 1302. In some embodiments, a 0 dB Gain Buffer 1306 iscoupled to the 4-Path filters 1301 to provide VBB,CH1. In someembodiments, a 0 dB Gain Buffer 1307 is coupled to the 4-Path filters1302 to provide V_(BB,CH2).

In some embodiments, the current I_(FB) passes through the switches S1of filters 1301 and is available at node Y (ignoring capacitance tosubstrate). In some embodiments, another correlator 1302 can be placedin series for concurrent reception of two signals. In some embodiments,the correlators 1301 and 1302 in each channel are driven by different PNsequences, PN_(CH1) and PN_(CH2). Impedance translation leads to aninput-match for the desired signals at LO frequency spread usingPN_(CH1) and PN_(CH2) while other signals are rejected.

In some embodiments, two series correlators in the feedback pathincreases the effective switch resistance and hence presents trade-offsbetween loss and linearity. Rejection of undesired signals may requirean equivalent RF current to be sourced by the LNA (Low Noise Amplifier)output. In some embodiments, higher series resistance R2 leads to largervoltage swing at the LNA output, limiting linearity. In someembodiments, increasing size of switches S1 increases parasiticcapacitance and/or resistance that shunt RF current to substratedegrading high frequency operation. In one example, switch resistance of15Ω is selected, achieving operation up to 1.4 GHz and 13 dBm OOB(out-of-band) IIP3 (third order intercept point).

In one example, the LNA is based on the design in providingapproximately 20 dB gain up to 1.4 GHz, consuming 25 mW. Two four-pathcorrelators are implemented in the feedback path. IQ buffer stages,following the baseband correlators, drive loads (e.g., 50Ω loads) with−0 dB gain. In this example, bandwidth of the N-path correlators andbuffer stages are 5 MHz and 3.4 MHz respectively.

In some embodiments, PN code modulation of the LO signal is doneoff-chip. The PN-code chip rate (e.g., 93.75 Mcs) is significantly lowerthan RF frequency and hence an on-chip LO code modulation schemeconsumes less power (e.g., less than 1 mW) in simulation.

FIG. 14A illustrates die photo 1400 of an integrated circuit (IC) of 65nm CMOS N-path RX, in accordance with some embodiments. In this example,the IC occupies 0.31 mm² in a 65-nm CMOS.

FIG. 14B illustrates plot 1420 showing measured S11 parameters whenW_(CH1) and W_(CH2) in FIG. 13A are 4-phase NOP pulses (without codemodulation) at two frequencies, f1 and f2. Here, f₂ is held constant at1 GHz while f₁ is varied. Here, while harmonic down conversion in N-pathmixers precludes harmonic relationship between f₁ and f₂, concurrentLO-tunable dual-frequency match can be observed.

FIG. 14C illustrates plot 1430 showing measured available and reflectedpower for matched and mismatched de-spreading codes, in accordance withsome embodiments. Here, an indirect code-domain matching measurementshown by waveforms 1431, 1432, and 1433 is performed using thecirculator-based setup.

FIG. 15A illustrates plot 1500 showing measured RX gain and isolationbetween a first channel and a second channel, in accordance with someembodiments. The measured spectrum available at RX port 2 is compared tothe measured spectrum at port 3 when de-spreading code in the RX is thesame and is different from the spreading code for the input signal. The10-dB higher reflected power when codes are different demonstratessignal reflection at the RF port, implying signal rejection at RF input.

The RX gain 1501 and 1502 for channels CH1 and CH2, respectively, ismeasured by applying 4-phase NOP pulses at 1 GHz and 0.6 GHz (FIG. 15A),with channels CH1 and CH2 achieving 35.5 dB and 38.5 dB gain,respectively (asymmetry due to frequency/position in feedback path).Leakage signal 1503 and 1504 from channels CH2 to CH1, respectively, ismeasured as the channel CH1 output with an RF input close to channel CH2LO frequency. Measurements demonstrate 35 dB isolation. Measured RX NF(noise figure) is 2.5 dB to 4 dB (e.g., 0.3 MHz to 1.4 GHz) withmeasured −26 dBm in-band and +13 dBm OOB IIP3 (for 45 MHz and 91 MHzoffset tones).

FIG. 15B illustrates plot 1520 showing measured two-tone SI at RX outputfollowing spreading and de-spreading, in accordance with someembodiments. Here, two waveforms 1521 and 1522 are shown. TX SIRC for atwo-tone SI at 750 MHz that has been spread using PN_(WF,1) is shown inFIG. 15B. De-spreading with PN_(X) provides SI rejection as indicated by1522. However, de-spreading with PN_(WF,2) as shown by 1521 demonstratessignal translation to 46.9 MHz for chip rate 93.75 Mcs. In someembodiments, subsequent low-pass filtering can provide higher SIrejection for PN_(WF,2) case.

FIG. 15C illustrates plot 1530 showing measured integrated in-band RXoutput power for modulated SI, in accordance with some embodiments.Here, integrated in-band power 1523 at RX baseband is shown as afunction of input power of 1.46 Mb/s QPSK modulated SI with 1 MHzbaseband integration bandwidth. Measurements demonstrate −11.8 dBminput-referred P_(1 dB) with respect to SI (translating to 13.2 dBm TXoutput assuming 25 dB TX/RX isolation). Here, the RX provides 35.5 dBgain for the desired signal, implying in-band SI rejection of 38.5 dB(from FIG. 15C) and +23.7 dBm OP_(1 dB) with respect to SI.

FIG. 15D shows the measured constellation 1540 for two concurrent 750MHz 1.46 Mb/s QPSK signals spread with different PN codes at 93.75 Mcsdemonstrating concurrent reception. Input power is limited by theinput-referred oscilloscope noise.

FIG. 15E shows the RX constellation recovery 1550 with SIRC for −40 dBmdesired input and −13.5 dBm SI (both at 1.46 Mb/s QPSK).

The code-domain approach of some embodiments is compared to prior SIRCapproaches in Table 1.

TABLE 1 Yang, Broek, Zhou, This work JSSC 15 [7] ISSCC 15 ISSCC 16 f_(c)(GHz) 0.3-1.4 0.5-1.5 0.8-1.4 0.15-3.5 0.6-0.8 Arch Code-domain N-Mixer-First Freq. Dom Mixer First Circulator+BB path Mixer TX/RX Eq. VMSI SIC No. of RX O/P 2 1 1 1 1 RX Gain (dB) 35/38 53 27-42 24 42 RX NF2.5-4 + 8 4/8 6.3 5.0 (dB) *0.9 dB filter mismatch SIRC NF Deg 0 dB —1.1 dB 4-6 dB 6.5 dB* IB IIP3 (dBm) −26 −38.7 −20 8/16.2 −33 Sl iPdB dB)@Gain (dB) −11.8 −9.7** −7.7** 9.3** −27.7** 35 53 27 24 42 SIRC 38.5 dB33 dB 25 dB 27 dB 42 dB BW 1 MHz 600 KHz 20 MHz 16.25 MHz 12 MHz* Power25.5 mW LNa 56 mW 250 mW 23 to 199 mW* 9.5 mW LO (incl. 56 mW ~1 mW codemod. TX BB) (sim) Area (mm²) 03.1 1.5 4.8 2 1.4 *includes circulator,**computed from measured IIP3 (this work measured integrated power)

The RX of some embodiments is the first code-domain N-path RX for SIRCand achieves low power consumption, smaller area and low NF degradation,showing a path towards combining code-domain and SIC techniques forSTAR.

FIG. 16 illustrates flowchart 1600 of a method of filtering, inaccordance with some embodiments. While various blocks shown here arearranged in a particular order, the order is not fixed. For example,some blocks may be executed before others while some may be executed inparallel to other blocks.

At block 1601, an RF input is received by a receiver via an array ofantennas. At block 1602, the RF input is provided to the N-path filter,wherein the N-path filter is coupled to the array of antennas. At block1603, a combination of non-overlapping pulses and a pseudo noise (PN)code is applied to the RF input. In some embodiments, the N-path filteris a spatio-spectral notch filter, and wherein the method comprisesconcurrently rejecting two blockers at two independent frequencies orangle-of-incidence at each input of an antenna of the array. In someembodiments, the N-path filter comprises switches coupled to at leastone antenna of the array. In some embodiments, at block 1604, theswitches are controlled by a code sequence generated by a combination ofthe non-overlapping pulses, Walsh-functions, and the PN code. In someembodiments, at block 1605, a local oscillating signal is converted intothe non-overlapping pulses as discussed with various figures.

Elements of embodiments (e.g., flowchart 1600 and scheme described withreference to FIGS. 1-15) are also provided as a machine-readable medium(e.g., memory) for storing the computer-executable instructions (e.g.,instructions to implement any other processes discussed herein). Themachine-readable medium (e.g., memory) may include, but is not limitedto, flash memory, optical disks, CD-ROMs, DVD ROMs, RAMs, EPROMs,EEPROMs, magnetic or optical cards, phase change memory (PCM), or othertypes of machine-readable media suitable for storing electronic orcomputer-executable instructions. For example, embodiments of thedisclosure may be downloaded as a computer program (e.g., BIOS) whichmay be transferred from a remote computer (e.g., a server) to arequesting computer (e.g., a client) by way of data signals via acommunication link (e.g., a modem or network connection).

Program software code/instructions associated with flowchart 1600(and/or various embodiments) and executed to implement embodiments ofthe disclosed subject matter may be implemented as part of an operatingsystem or a specific application, component, program, object, module,routine, or other sequence of instructions or organization of sequencesof instructions referred to as “program software code/instructions,”“operating system program software code/instructions,” “applicationprogram software code/instructions,” or simply “software” or firmwareembedded in processor.

Reference in the specification to “an embodiment,” “one embodiment,”“some embodiments,” or “other embodiments” means that a particularfeature, structure, or characteristic described in connection with theembodiments is included in at least some embodiments, but notnecessarily all embodiments. The various appearances of “an embodiment,”“one embodiment,” or “some embodiments” are not necessarily allreferring to the same embodiments. If the specification states acomponent, feature, structure, or characteristic “may,” “might,” or“could” be included, that particular component, feature, structure, orcharacteristic is not required to be included. If the specification orclaim refers to “a” or “an” element, that does not mean there is onlyone of the elements. If the specification or claims refer to “anadditional” element, that does not preclude there being more than one ofthe additional element.

Furthermore, the particular features, structures, functions, orcharacteristics may be combined in any suitable manner in one or moreembodiments. For example, a first embodiment may be combined with asecond embodiment anywhere the particular features, structures,functions, or characteristics associated with the two embodiments arenot mutually exclusive.

The following examples are provided with reference to variousembodiments.

Example 1

An apparatus comprising: a receiver; and one or more N-path filterscoupled to an input of the receiver, wherein the one or more N-pathfilters apply a combination of non-overlapping pulses and a pseudo noise(PN) code.

Example 2

The apparatus of example 1 comprises an array of antennas coupled to oneor more N-path filters, wherein the array of antennas comprisemultiple-input-multiple-output (MIMO) array.

Example 3

The apparatus of example 2, wherein the one or more N-path filters areused as one or more spatio-spectral notch filters which are toconcurrently reject two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.

Example 4

The apparatus of example 2, wherein the one or more N-path filterscomprise switches coupled to at least one antenna of the array.

Example 5

The apparatus of example 4, wherein the switches are controllable by acode sequence generated by a combination of the non-overlapping pulses,Walsh-functions, or the PN code.

Example 6

The apparatus of example 5 comprises circuitry to convert a localoscillating signal into the non-overlapping pulses.

Example 7

The apparatus of example 5, wherein at least one switch of the switchesis coupled to a capacitor and a resistor, wherein the resistor iscoupled in parallel to the capacitor.

Example 8

The apparatus of example 5, wherein at least one switch of the switchesis coupled to an inductor and a resistor, wherein the resistor iscoupled in parallel to the inductor.

Example 9

The apparatus of example 1, wherein receiver comprises a code-domainN-path receiver.

Example 10

The apparatus of example 1, wherein the PN code includes one of: Goldcode, Kasami code, Barker code, or M-sequences.

Example 11

A method comprising: receiving an RF input by a receiver via an array ofantennas; providing the RF input to one or more N-path filters; andapplying a combination of non-overlapping pulses and a pseudo noise (PN)code to the RF input.

Example 12

The method of example 11, wherein the one or more N-path filters are oneor more spatio-spectral notch filters, and wherein the method comprisesconcurrently rejecting two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.

Example 13

The method of example 11, wherein the one or more N-path filterscomprise switches coupled to at least one antenna of the array.

Example 14

The method of example 14 comprises controlling switches by a codesequence generated by a combination of the non-overlapping pulses,Walsh-functions, or the PN code.

Example 15

An apparatus comprising: an array of antennas that receives multiple RFinputs and provides multiple radio frequency (RF) or intermediatereference (IF) outputs; and one or more N-path filters to receive the RFor IF outputs, wherein the one or more N-path filters include switchesfor applying a combination of non-overlapping pulses and a pseudo noise(PN) code to the RF or IF outputs.

Example 16

The apparatus of example 15, wherein the one or more N-path filterscomprise one or more spatio-spectral notch filters.

Example 17

The apparatus of example 16, wherein the one or more N-path filters areto concurrently reject two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.

Example 18

The apparatus of example 15, wherein switches of the one or more N-pathfilters are coupled to at least one antenna of the array.

Example 19

The apparatus of example 15 comprises circuitry to control the switchesby a code sequence generated by a combination of the non-overlappingpulses, Walsh-functions, or the PN code.

Example 20

The apparatus of example 15, wherein at least one switch of the switchesis coupled to a capacitor and a resistor, wherein the resistor iscoupled in parallel to the capacitor.

While the disclosure has been described in conjunction with specificembodiments thereof, many alternatives, modifications and variations ofsuch embodiments will be apparent to those of ordinary skill in the artin light of the foregoing description. The embodiments of the disclosureare intended to embrace all such alternatives, modifications, andvariations as to fall within the broad scope of the appended claims.

In addition, well known power/ground connections to integrated circuit(IC) chips and other components may or may not be shown within thepresented figures, for simplicity of illustration and discussion, and soas not to obscure the disclosure. Further, arrangements may be shown inblock diagram form in order to avoid obscuring the disclosure, and alsoin view of the fact that specifics with respect to implementation ofsuch block diagram arrangements are highly dependent upon the platformwithin which the present disclosure is to be implemented (i.e., suchspecifics should be well within purview of one skilled in the art).Where specific details (e.g., circuits) are set forth in order todescribe example embodiments of the disclosure, it should be apparent toone skilled in the art that the disclosure can be practiced without, orwith variation of, these specific details. The description is thus to beregarded as illustrative instead of limiting.

We claim:
 1. An apparatus comprising: a receiver, which comprises acode-domain N-path receiver; one or more N-path filters coupled to aninput of the receiver, wherein the one or more N-path filters apply acombination of non-overlapping pulses and a pseudo noise (PN) code; andan array of antennas coupled to one or more N-path filters, wherein thearray of antennas comprises multiple-input-multiple-output (MIMO) array.2. The apparatus of claim 1, wherein the one or more N-path filtersoperate as one or more spatio-spectral notch filters which are toconcurrently reject two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.
 3. The apparatus of claim 1, wherein the one or more N-pathfilters comprise switches coupled to at least one antenna of the array.4. The apparatus of claim 3, wherein the switches are controllable by acode sequence generated by a combination of the non-overlapping pulses,Walsh-functions, or the PN code.
 5. The apparatus of claim 4 comprisescircuitry to convert a local oscillating signal into the non-overlappingpulses.
 6. The apparatus of claim 4, wherein at least one switch of theswitches is coupled to a capacitor and a resistor, wherein the resistoris coupled in parallel to the capacitor.
 7. The apparatus of claim 4,wherein at least one switch of the switches is coupled to an inductorand a resistor, wherein the resistor is coupled in parallel to theinductor.
 8. The apparatus of claim 1, wherein the PN code includes oneof: Gold code, Kasami code, Barker code, or M-sequences.
 9. A methodcomprising: receiving a radio frequency (RF) input by a receiver via anarray of antennas; providing the RF input to one or more N-path filters,wherein the one or more N-path filters comprise switches coupled to atleast one antenna of the array; and applying a combination ofnon-overlapping pulses and a pseudo noise (PN) code to the RF input. 10.The method of claim 9, wherein the one or more N-path filters are one ormore spatio-spectral notch filters, and wherein the method comprisesconcurrently rejecting two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.
 11. The method of claim 9 comprises controlling the switches by acode sequence generated by a combination of the non-overlapping pulses,Walsh-functions, or the PN code.
 12. An apparatus comprising: an arrayof antennas that receives multiple RF inputs and provides multiple radiofrequency (RF) or intermediate reference (IF) outputs; and one or moreN-path filters to receive the RF or IF outputs, wherein the one or moreN-path filters include switches for applying a combination ofnon-overlapping pulses and a pseudo noise (PN) code to the RF or IFoutputs.
 13. The apparatus of claim 12, wherein the one or more N-pathfilters comprise one or more spatio-spectral notch filters.
 14. Theapparatus of claim 13, wherein the one or more N-path filters are toconcurrently reject two or more blockers at two or more independentfrequencies or angle-of-incidence at each input of an antenna of thearray.
 15. The apparatus of claim 12, wherein the switches of the one ormore N-path filters are coupled to at least one antenna of the array.16. The apparatus of claim 12 comprises circuitry to control theswitches by a code sequence generated by a combination of thenon-overlapping pulses, Walsh-functions, or the PN code.
 17. Theapparatus of claim 12, wherein at least one switch of the switches iscoupled to a capacitor and a resistor, and wherein the resistor iscoupled in parallel to the capacitor.
 18. An apparatus comprising: areceiver; and one or more N-path filters coupled to an input of thereceiver, wherein the one or more N-path filters apply a combination ofnon-overlapping pulses and a pseudo noise (PN) code, and wherein the oneor more N-path filters comprise switches coupled to at least one antennaof an array of antennas.
 19. The apparatus of claim 18, wherein theswitches are controllable by a code sequence generated by a combinationof the non-overlapping pulses, Walsh-functions, or the PN code.
 20. Theapparatus of claim 19, comprises circuitry to convert a localoscillating signal into the non-overlapping pulses.
 21. The apparatus ofclaim 18 comprises the array of antennas coupled to the one or moreN-path filters, wherein the array of antennas comprisesmultiple-input-multiple-output (MIMO) array.